[time-nuts] On low-voltage TAC/TDCs for a GPSDO

Bruce Griffiths bruce.griffiths at xtra.co.nz
Fri Aug 13 05:59:51 UTC 2010


Yet another option is to sample the output of a simple 1us time constant 
RC low pass filter and fit an exponential to the sampled data and 
calculate the threshold crossing from this.
If the aberrations are sufficiently low over the range of time intervals 
measured (0.5us to 1us with a conventional 2 stage synchroniser clocked 
at 2MHz) this would be the simplest solution in that it only requires a 
single resistor and a single capacitor.

Bruce


Bruce Griffiths wrote:
> Yes, with a 2MSPS ADC and 1-2us transition times one gets 2-4 samples 
> during the transition.
> Worst case with a 1us filter (10%-90%) output transition time there 
> may be one sample at the midpoint and samples close to the 10% and 90% 
> amplitude points.
> 2us transition times are probably close to optimum.
> In the latter case the effective time stamp resolution (with a true 12 
> bit ADC) will be around 0.5ns.
>
> Ideally a gaussian impulse response filter should be used.
> However if the input transitions are sufficiently (to allow the filter 
> transients to settle) far apart almost any reasonable (without 
> excessive overshoot) could be used.
> The minicircuits LPF_BOR3+ low pass filter appears almost good enough.
>
> Bruce
>
> Bob Camp wrote:
>> Hi
>>
>> Would't you want 2 or more samples during the transition?
>>
>> Bob
>>
>>
>>
>> On Aug 12, 2010, at 8:25 PM, Bruce 
>> Griffiths<bruce.griffiths at xtra.co.nz>  wrote:
>>
>>> Another method is to attenuate (to within the ADC input range) the 
>>> PPS signal to be timestamped, low pass filter it and capture a 2MSPS 
>>> sample burst centred around the low pass filter output transition 
>>> midpoint.
>>> You can then use WKS interpolation to time stamp the transition 
>>> midpoint (when it crosses a threshold halfway between the initial 
>>> and final values of the low pass filter output).
>>> The low pass filter (preferably an LC filter) delay is easily 
>>> calibrated by timestamping an internally generated signal initiated 
>>> on a known ADC sampling clock edge.
>>> No (external) current sources, reset switches etc are required.
>>> With a 2MSPS sample rate a low pass filter output transition time of 
>>> 1-2us should suffice (provided the ADC has a sufficiently large 
>>> large signal bandwidth).
>>>
>>> Bruce
>>>
>>> Bruce Griffiths wrote:
>>>> Some options:
>>>>
>>>> 1) Use a 74AHC05 for Q1 and Q2.
>>>>
>>>> 2) Switch the current source at the emitter node and only turn on 
>>>> the current source when charging the capacitor.
>>>> This will increase the available TAC output voltage range and/or 
>>>> improve the linearity by eliminating the diode.
>>>> However the capacitor discharge switch should be turned off before 
>>>> charging the capacitor.
>>>> A stable fixed delay of a few (10ns??) before switching on the 
>>>> current source is required.
>>>>
>>>> 3) Replace the current source with a resistor.
>>>> The resultant nonlinearity is well defined and software correction 
>>>> should be relatively easy.
>>>>
>>>> 4) If the ADC(s) have a sufficiently wide full power bandwidth then 
>>>> one could just sample a pair of quadrature phased 250kHz sinewaves.
>>>> Extend the range by sampling (synchronise the input sampling edge 
>>>> to the counter clock first) a counter clocked at 250KHz.
>>>> Initiate the sampling with the signal edge to be time stamped.
>>>>
>>>> If the GPSDO is used to clock the microprocessor, counters and 
>>>> produce the quadrature sinewave outputs then only a single TDC 
>>>> (time to digital converter) is required.
>>>>
>>>> Measuring negative time intervals should not be necessary as the 
>>>> TAC (or other TDC) should be used merely to measure the delay of a 
>>>> synchroniser the output of which is used to synchronously sample a 
>>>> counter clocked with the same clock as the synchroniser.
>>>>
>>>>
>>>> J.D. Bakker wrote:
>>>>> Hello all,
>>>>>
>>>>> I'm working on Yet Another DIY GPSDO, and one of the issues I've 
>>>>> been looking into is a TAC/TDC to do sawtooth correction on the 
>>>>> measurement of the GPS PPS signal. I'd like to stick with a 3.3V 
>>>>> supply for most of the circuit, and several of the TAC designs 
>>>>> that have been discussed here in the past run into trouble at such 
>>>>> low voltages (mostly through VBE drops).
>>>>>
>>>>> To start with the context: I'm planning to use a microcontroller 
>>>>> with a built-in dual 12-bit 2MSPS ADC. I'd like to not use 
>>>>> anything that's not available at Digi-Key or Mouser, and keep the 
>>>>> SMD pitch>=0.8mm (with a possible exception for dual transistors 
>>>>> in SOT-23-6). That way the design shouldn't be too hard for others 
>>>>> to replicate.
>>>>>
>>>>> I'm aiming for a TAC accuracy of 1ns, allowing for one or a few 
>>>>> calibrations between PPS pulses. Minimum full-scale range should 
>>>>> be +/- a few hundred ns, to allow for outliers. (The plan is to 
>>>>> have an initial FLL for coarse locking, and have the PLL kick in 
>>>>> after that). I'm penciling in an ADC reference voltage of 2V, as 
>>>>> that's commonly available and leaves enough headroom to use the 
>>>>> current sources in their most linear range.
>>>>>
>>>>> I've attached a diagram that reflects a few of my current thoughts.
>>>>>
>>>>> - Circuit 1 is the traditional TAC. Before the start of the cycle 
>>>>> Q2 conducts, discharging C1 and shunting I1's current to ground. 
>>>>> At this point the ADC can measure the voltage drop across C1/Q2 to 
>>>>> eliminate that offset. Taking nSTART low puts Q2 into 
>>>>> high-impedance, and I1 charges C1 through D1 until STOP is raised 
>>>>> causing Q1 to shunt I1's current to ground. At this point the ADC 
>>>>> samples the voltage across C1, which is proportional to the time 
>>>>> between START and STOP (modulo offset and nonlinearities).
>>>>>
>>>>> This circuit is well known to work (although it is more common to 
>>>>> use Q1 for both START and STOP and to limit Q2 to ramp discharge 
>>>>> duties). Downsides are that negative time offsets cannot be 
>>>>> measured directly, and the constant output voltage offers little 
>>>>> room for increased precision through sample averaging, unless the 
>>>>> ADC's input noise is large compared to its LSB size. For the same 
>>>>> reason there is no easy way to reduce the effects of ADC INL/DNL.
>>>>>
>>>>> - Circuit 2 works in a similar way, except that the ramp isn't 
>>>>> terminated by a STOP signal but is allowed to run freely until I2 
>>>>> saturates. The ADC is set to sample continuously, taking multiple 
>>>>> samples of the ramp, and the microcontroller interpolates the 
>>>>> resulting values to determine the elapsed time between an internal 
>>>>> time reference point and the START signal.
>>>>>
>>>>> This circuit is fairly simple, and has the advantage that there is 
>>>>> no hard limit to its range. Curve-fitting the sampled values 
>>>>> increases precision and reduces the effects of INL/DNL. On the 
>>>>> other hand, ADC aperture jitter and offset have a direct impact on 
>>>>> resolution.
>>>>>
>>>>> - Circuit 3 expands on this approach by having dual ramp 
>>>>> generators, and having the ADC measure the voltage difference 
>>>>> between the two.
>>>>>
>>>> Not a good idea, as this requires accurate matching of the gains of 
>>>> the 2 TACs.
>>>> Its better to sample each TAC output individually as this allows 
>>>> software correction for gain mismatch (and nonlinearity) before 
>>>> subtraction.
>>>> Software correction is better than using trimpots or similar as the 
>>>> parasitics etc associated with trimpots are eliminated.
>>>>
>>>>> This approach is the only one of the three that can directly 
>>>>> measure negative time offsets, allowing a regenerated pulse to be 
>>>>> directly compared with the GPS' PPS. A small difference in ramp 
>>>>> rates, unavoidable in practice, actually helps to average out DNL 
>>>>> and is easily corrected in calibration. Sampling time uncertanties 
>>>>> have less impact than in Circuit 2. Then again, it may be 
>>>>> difficult to reliably detect the start/end-of-ramp points from the 
>>>>> samples alone. Total range is relatively limited, and due to the 
>>>>> differential measurements it is harder to reduce current source 
>>>>> nonlinearities in software.
>>>>>
>>>>> Any thoughts? At this point I'm tempted to build a hybrid of 2 and 
>>>>> 3, using one of the microcontroller's ADCs in each mode.
>>>>>
>>>>> I've not seen prior work on the ramp-approach, although it's a 
>>>>> close cousin to the centroid pulse timing method 
>>>>> (<http://www.febo.com/pipermail/time-nuts/2006-September/021765.html>). 
>>>>> Has anyone seen it before (and possibly shot down due to major 
>>>>> deficiencies)? It seems too obvious to not have been considered by 
>>>>> others.
>>>>>
>>>> It was usually not feasible as the ADC's typically used had 
>>>> insufficient input power bandwidth.
>>>> The settling time and power bandwidth of any buffer amplifier 
>>>> between the ramp capacitor and the ADC has also to be considered.
>>>> If one uses a capacitive input charge redistribution ADC connected 
>>>> directly to the ramp capacitor then the sampling process itself 
>>>> transfers charge from the ramp capacitor to the sampling capacitor. 
>>>> Software compensation for this effect may be required as the 
>>>> transferred charge depends on the number of samples from the ramp 
>>>> start to the current sample.
>>>> You will also need to ensure that the current source recovers 
>>>> sufficiently quickly from saturation.
>>>>
>>>> Another issue is to limit the discharge current flowing in the 
>>>> discharge switch.
>>>> Often a 2 step discharge is used.
>>>> A switch with a series resistor is used to discharge the capacitor 
>>>> to the point at which the second switch can be turned on to 
>>>> complete the discharge without excessive curent flowing in this 
>>>> switch. See the HP53131A/2A schematics for an example.
>>>>
>>>>> (Notes: These are initial rough sketches. The ramp current has not 
>>>>> been optimized yet; I have an unsubstantiated feeling that 
>>>>> brute-forcing it with a higher current and larger cap may well 
>>>>> help to swamp some of the nonlinearities. I've mostly picked 1V/us 
>>>>> ramp speed out of the air because it gives me 4-5 samples @2MSPS 
>>>>> which is a workable number to do curve fitting on. Also not sure 
>>>>> whether I'll use the simpler one-transistor current source or the 
>>>>> hi-Zout mirror with a current source derived from the ADC's 
>>>>> reference. The FETs may end up being implemented as single-gate 
>>>>> /OE drivers. I'll do a more complete write-up on the entire GPSDO 
>>>>> later).
>>>>>
>>>>> Thanks,
>>>>>
>>>>> JDB.
>>>>>
>>>> Bruce
>>>>
>>>>
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