[time-nuts] Sub Pico Second Phase logger

Bruce Griffiths bruce.griffiths at xtra.co.nz
Wed Dec 10 02:20:08 UTC 2008


Joe

Joseph M Gwinn wrote:
> Bruce,
>   
>>> By the way, despite the circuit diagram in the datasheet, the 
>>> corresponding phase-detector module MPD-1 can be wired to have the IF 
>>> output ground isolated from the common RF, LO and case ground.  A 
>>> little work with an ohmmeter will tell the tale.  This can help to 
>>>       
> contain the 
>   
>>> low frequency beatnote.
>>>       
>> Yes, that's usually the case for the Minicircuits PCB mount phase
>> detectors and mixers except for some surface mount versions (usually the
>> very high frequency models).
>>     
>
> So it was already known.  It looks to me that MiniCircuit's intent is to 
> support automated testing of modules.
>
>
>   
>> A PCB mount mixer package is also preferable as its then much easier to
>> use a capacitive IF port termination (for lower noise) in conjunction
>> with series resistors at the RF and LO ports (for lower VSWR) than if a
>> mixer with SMA or other coax connectors were used.
>>     
>
> I've been using 3 and 8 dB coaxial attenuators at the LO and RF inputs 
> respectively, and it makes a big difference.
>
> But I don't understand the part about capacitive loading of the IF port. I 
> would think that the low pass filter would need to present a matched 
> impedance at the sum frequency, so the emerging high-level 20 MHz signal 
> is not reflected back into the mixer.
>   
Reflecting the sum frequency back into the mixer is actually necessary
to reduce the noise at the IF port.
I believe that one of Agilent's simulation application notes mentions
this effect but I don't recall the actual application note number.
This will affect the mixer RF and IF port impedance so adding a series
resistor may be required to improve the SWR.
> MiniCircuits AN-41-001 "FAQ about Phase Detectors" has on page 2 a 500 ohm 
> resistor to ground and a 5000 ohm resistor to the first filter capacitor, 
> so the capacitor is isolated from the IF port by the resistors.
>
>   
I wouldn't take too much notice of that recommendation as I have little
confidence in the author's experience/knowledge.

>> Supposedly an SRA-1, but some caution is in order as some statements as
>> to the effect of the input offset of an opamp based IF preamp in the
>> same application note were of dubious veracity unless one were to use an
>> inverting opamp input stage.
>>     
>
> This issue was mentioned in another app note, but their main issue 
> appeared to be that the opamp bias currents could cause an offset.
>
>
>   
But the circuit they suggest has no effect on bias current induced
offset, the same current flows into the mixer and termination impedance
independent of the series resistance.
>>>> A NIST paper indicated that mixer phase shift tempco was around 10x
>>>> lower if the RF port was unsaturated. It also indicated that the 
>>>>         
> mixer
>   
>>>> phase shift tempco is much lower if the input frequency is 100MHz 
>>>>         
> rather
>   
>>>> than 10MHz. This was one reason given for shifting to 100MHz 
>>>> DMTD systems.
>>>>
>>>>         
>>> Do you recall which paper?
>>>
>>>
>>>
>>>       
>> http://tf.nist.gov/timefreq/general/pdf/971.pdf
>> <http://tf.nist.gov/timefreq/general/pdf/971.pdf>
>> Has some measurement data on mixer phase shift tempco and power
>> sensitivity and their frequency dependence etc.
>>     
>
> I do know this paper.  At the bottom of page 834, to the right, is the 
> estimate 3.5 pS/K.
>
> Another reason to go to 100 MHz is that the temperature coefficient of 
> electrical length of polymer-insulated coax is far lower at 100 MHz 
> compared to 10 MHz.
>
>
>   
>> I'll search for the paper that stated that the phase shift tempco was
>> lower if the RF port was unsaturated.
>>     
>
> I think I have seen this too, but don't recall where.  But it's why I use 
> an 8 dB pad on the RF input.
>
>
> http://www.wj.com/archive/documents/Tech_Notes_Archived/Mixers_phase_detectors.pdf
>   
>> . 
>>     
>>> Don't know how long this URL will work, as WJ is assimilated into 
>>> TriQuint.
>>>       
>
> This app note is reference 7 of paper 971 above.
>
>
> [Soundcards]
>>>
>>>> With identical beat frequency outputs, crosstalk between 
>>>> channels within the sound card shouldn't be a great problem.
>>>>
>>>>         
>>> I'm not sure I believe this, as there is likely ground coupling within 
>>>       
> the 
>   
>>> soundcard and the ear is famously insensitive to phase.  Channel 
>>>       
> isolation 
>   
>>> of 60 dB isn't enough to prevent phase shifts.
>>>
>>>
>>>       
>> It will be present but its effect in some cases (when the phase shift
>> between channels is such that the crosstalk phase is at 90 degrees to
>> the signal of interest) will be negligible, in other cases it is easily
>> measured and compensated for.
>>     
>
> Isn't 90 degrees (quadrature) the worst case for causing phase shifts? 
>
>   
Yes, I should have said that when the 2 input signals are in quadrature,
any capacitive crosstalk will have little effect on the phase shift.
> To get a one picosecond change at 10 MHz by injection of an attenuated 
> quadrature copy of the main signal requires a relative voltage ratio of 
> Tan[(10^-12)(10^7)(360)] = Tan[0.0036] = 0.0000628 of the main signal, or 
> 20 Log[0.0000628]= -84 dBc.  This is well exceeds the interchannel 
> isolation of many sound cards.
>
> Cancellation by mathematical means could be possible, but will require a 
> dynamic range well exceeding 84 dB.  This ought to be easy to arrange.
>
>   
The AP192 has a somewhat higher interchannel isolation than that, the
interchannel crosstalk spec is about -120dB.
With a sufficiently large number of samples the its easy to see
artifacts as low as -140dBFS.
>   
>>>> One concern particularly for low beat frequencies is the phase shift 
>>>>         
> in
>   
>>>> the sound card input coupling capacitors (usually electrolytics).
>>>>
>>>> It should be easy to test the sound card phase shift stability for 
>>>>         
> this
>   
>>>> application by driving both inputs from the same signal source.
>>>>         
>
> Or terminate one channel input and drive the other, and measure the 
> amplitude and phase of whatever comes out of the terminated channel, 
> compared to the driven channel.  Then swap channels and repeat.  The phase 
> and amplitude will depend on frequency, so a sweep will be required, and 
> some frequencies may need to be avoided.
>
>
>   
>>>       
>> Joe
>>
>> I suspect that slow phase changes much less than 1ns or so are hard to
>> distinguish from gain drift given the gain tempco of the ECL 
>> phase detector.
>>
>> A beat note near 1kHz appears to be even better if one is using
>> something like an enhanced Costas receiver or even using WKS
>> interpolation to locate and time stamp zero crossings.
>>     
>
> But it limits the phase slope gain.  I suppose there is an optimum 
> somewhere.
>
>  
>   
>> So far only the M-Audio AP192 has been used.
>> Tests with an embedded motherboard 16 bit sound system show
>> significantly increased noise.
>> I've found that the noise level of motherboard sound systems varies
>> enormously from one motherboard model (sample of 2) to another.
>>
>> Any 24 bit sound card with a performance close to or better than that of
>> the AP192 should suffice.
>>     
>
> In general, firewire connected sound cards should be better, because the 
> soundcard maker has complete control of what's inside the box.  Unlike 
> inside a PC. 
>
>   
Its hard to find such Firewire systems without such unnecessary frills
(for this application) as high gain preamps.
The gain tempco and linearity of some variable gain audio preamps is
somewhat suspect.
>   
>> Other cards using AKM 24 bit ADCs should also be suitable.
>>     
>
> Who is AKM?
>
>   
Asahi Kasei EKM

http://www.asahi-kasei.co.jp/akm/en/

http://www.asahi-kasei.co.jp/akm/en/product/proaudio.html

> 20 Log[ 2^24 ] = 144 dB, so something else will be the limit.
>
>
>   
Actual ENOB ~ 19 to 20 bits.
>> Ideally an external sound card with balanced  XLR inputs would be best.
>>
>> HP produced a number of different phase comparators each with a
>> different type of phase detector.
>> The K34-5991A design can't be older than the early 1970's because the
>> MECLIII devices used weren't available until then.
>>     
>
> OK.  I recall MECL.  RIP.  But we have PECL now.
>
>   
Same thing, different supplies.
PECL when Vcc is +ve and Vee is GND.
NECL when Vcc is GND and Vee is -ve.
>   
>> Warren built a similar phase detector (differential XOR or XOR + XNOR)
>> using CMOS ICs and for a common 10MHz input with a phase difference near
>> zero found short term output noise of of around 10uV or so (10V phase
>> detector FSR) using a passive low pass filter.
>>     
>
> (10uV)/(10v)= 1 ppm.  (100 nS)(10^-6)= 0.1 pS.
>
>
>   
>> In principle an ADC like the LTC2484 could be used with a 2.5V CMOS
>> OR/XNOR phase detector and passive low pass filters.
>> The ratiometric conversion capability will significantly reduce the
>> sensitivity to the XOR gate supply if the XOR gate supply is also used
>> for the ADC reference voltage.
>> If one used 5V logic a resistive output attenuator would be needed which
>> reduce the gain stability somewhat.
>>
>> All such phase detectors suffer from substantial nonlinearity near the
>> ends of the range due to gate output slew rate limitations.
>>     
>
> If one is tracking through multiple phase cycles (as did the HP unit), 
> this would matter.
>
>   
Can alleviate it to some extent by driving a pair of such phase
detectors so that their outputs are in quadrature.
One just selects the phase detector output that is in the linear range.
The quadrature outputs also allow unambiguous assignment of the sign of
any phase change.
> Joe
>
>   

Bruce



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