[time-nuts] On low-voltage TAC/TDCs for a GPSDO
bruce.griffiths at xtra.co.nz
Sat Aug 14 11:49:50 UTC 2010
J.D. Bakker wrote:
> At 19:01 +1200 14-08-2010, Bruce Griffiths wrote:
>> J.D. Bakker wrote:
>>> At 08:30 +1200 14-08-2010, Bruce Griffiths wrote:
>>>> Using a synchroniser allows the TAC output range to be combined
>>>> with the coarse timestamp derived by sampling a counter clocked by
>>>> the same clock as the synchroniser.
>>> I think we're looking at it from two different angles.
>>> What I read from your description is close to the traditional
>>> architecture such as used in the HP5335A, with a counter running at
>>> the system clock frequency for coarse measurement and a TAC to
>>> measure the remainder. What I'm planning to do is more akin a
>>> traditional PLL, with the TAC as the Phase Detector. For this to
>>> work I assume that a coarse FLL (using a counter) has already
>>> brought the oscillator within lock range. Is there any reason that
>>> method won't work, or can trivially be made to work better?
>> Having a wide TAC range means that its resolution and noise depends
>> critically on that of the ADC.
>> Since some ADCs embeded within processor dont have true 12 bit
>> performance this may limit the TAC resolution/noise to several
>> nanosec rather than the desired 1ns or better.
> No, the TAC range would only be wide enough to cover the expected
> spread of valid PPS pulses from the GPS (say +/-500ns...+/-1us).
With some internal 12 bit ADCs that dont have true 12 bit you will
barely achieve 1ns resolution with a 2us range.
> (I've thought a bit more about what you proposed, ie using the TAC to
> measure synchronizer delay. Problem is I'd like to use the
> timestamping counter that's internal to the CPU, and I see no way of
> getting at the output of its built-in synchronizer. This could of
> course be fixed by using an external timestamping
> counter/synchronizer, but that seems like a bit of a waste of resources).
Surely you only need an external synchroniser (ie a dual D flipflop)
clocked by the same clock (or at least one synchronous with it) as the
The internal synchroniser then only adds a fixed delay.
>>> (The regenerated PPS output will indeed be derived from and
>>> synchronous with the VCXO/OCXO. It is also my intention to have the
>>> OCXO clock the microcontroller, either directly or through a
>>> prescaler, depending on whether the XO runs higher or lower than the
>>> max CPU clock).
>> That ensures that all intermod products are harmonically or
>> submultiples of the OCXO frequency.
> Indeed. I prefer knowing where my birdies are (and preferably placing
> them where they do the least harm), rather than having them drift over
> time, frequency and temperature.
>>>> The output compliance of your four transistor current mirror is
>>>> limited to around 1.3V or so before the onset of saturation or
>>>> gross nonlinearity.
>>> It's actually better than that, from what I can see from simulations
>>> and measurements. If the transistor currents are close to equal and
>>> the ramp rate isn't too high, output current stays within 1% up to
>>> ~1V, and the mirror saturates at 0.6-0.7V. This is with common
>>> small-signal transistors with an fT of a few hundred MHz.
>> There are 2xVbe + 1x diode drop to subtract from 3.3V ie somewhere
>> from 1.8V -2.4V leaving a ramp amplitude of 1.5V to 1.1V depending on
>> temperature and transistor current.
> That's what I thought when I first saw it and started counting
> junctions, but it's actually quite a bit better than that as the
> cross-coupling of the transistors steers current from saturating
> transistors into the bases of the opposing CE transistor. I found it
> in Barrie Gilbert's chapter on Bipolar Current Mirrors in the book
> "Analogue IC Design: the current-mode approach"; Google Books has a
> preview of much of this chapter.
Simulation appears to indicate otherwise, distortion starts to rise as
one of the mirror transistors nears saturation.
One way to look at this is to look at variations in ramp charging current.
However the ultimate test (other than breadboarding it) is to actually
simulate the sampling process and look at the deviation of the sampled
voltages from linearity.
In the case of the 3 diode TAC devised by Kasper Pedersen some
compensation of diode capacitance modulation occurs if the diodes are
> I've tried it in the simulator and on the bench, and it works quite
> well. If you want to test it I suggest increasing the current source
> to 10mA, the cap to 10nF and starting with 150R for R1/R2 plus 10R
> emitter resistors for the CE transistors. I've tested it with the
> common European BC5xx/BC8xx-types, but LTSpice seems to like it with
> 2N3906s too. In that configuration, the ramp stays within +/-150uV of
> a linear approximation over a ramp range between 0 and 2V when ramping
> at 1V/us, which corresponds to +/-0.6LSB for a 12-bit ADC.
I'll check again, but thats not consistent with what I found with a
simulated 1mA current source.
The capacitor charging current started to deviate significantly as
saturation was approached.
I also simulated other current sources with higher output resistance and
somewhat greater compliance.
>>> [I should probably make a sketch of the entire GPSDO and post it]
>> Yes that would be useful as details can often be important.
> I have to leave now, will do so when I get back.
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